Multi-phase buck converter

ABSTRACT

A buck converter for providing an output voltage to a load, including a plurality of output switch arrangements having respective output inductors coupled to an output capacitor, the switch arrangements being controllable to provide phase output currents to the output capacitor; a plurality of phase output arrangements respectively coupled to the output switch arrangements, the phase output arrangements being controllable to set the respective phase output currents supplied by the output switch arrangements; a phase control bus communicatively coupled to each of the phase output arrangements; and a phase control arrangement communicatively coupled to the phase control bus, the phase control arrangement being configured to control the phase output arrangements to set the respective phase output currents so that the output voltage approximates the desired voltage; wherein the phase control arrangement and the phase output arrangements are provided as respective integrated circuits.

RELATED APPLICATIONS

The present application is a divisional application of U.S. patentapplication Ser. No. 10/914,707, filed Aug. 9, 2004, which is acontinuation application of U.S. patent application Ser. No. 10/392,121,filed Mar. 18,2003, entitled MULTI-PHASE BUCK CONVERTER, whichapplication is based on and claims the benefit of U.S. ProvisionalApplication No. 60/366,889, filed on Mar. 22, 2002, entitled SYNCHRONOUSBUCK CONVERTER WITH MULTIPLE PHASES, the entire contents of which areexpressly incorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to buck converters, such as multi-phasebuck converters for use in low voltage/high-current applications.

BACKGROUND INFORMATION

Various applications may provide a conventional DC-to-DC buck converterthat accepts a DC input voltage and produces a lower DC output voltageto drive at least one circuit component. Buck converters are typicallyused in low voltage applications requiring high amounts of load current(e.g., 30 amps or more). Typically, as shown in FIG. 19, a single phasebuck converter 1900 includes a high-side switch 1905, a low-side switch1910 connected to the high-side switch at a switch node 1915, an outputinductor 1920 connected to the switch node 1915, and an output capacitor1925 connected to the output inductor 1920.

In operation, the high-side and low-side switches 1905, 1910 arecontrolled by a control circuit 1930 to produce a desired output voltageacross a load 1935. For this purpose, the high-side switch 1905 isinitially switched on, while the low-side switch 1910 remains off. Thiscauses a voltage drop across the output inductor 1920 of approximately(V_(IN)−V_(OUT)), which causes a current to build inside the outputinductor 1920. At a subsequent time, the high-side switch 1905 isswitched off, and the low-side switch 1910 is switched on. Since thecurrent within the inductor 1920 cannot change instantly, sourcedthrough switch 1910, the current continues to flow through the outputinductor 1920, thereby charging the output capacitor 1925 and causingthe voltage (V_(OUT)) across the output capacitor 1925 to rise.

In this manner, the high-side and the low-side switches 1905, 1910 maybe suitably switched at appropriate times, until the voltage (V_(OUT))across the output capacitor 1925 equals a desired output voltage, whichis typically lower than the input voltage. Once the desired outputvoltage is reached, the high-side and the low-side switches 1905, 1910may be periodically controlled so that the output inductor 1920 providesan amount of current equal to the current demand of a load 1935connected across the output capacitor 1925. By providing no more and noless than the current demand of the load 1935, the voltage (V_(OUT))across the output capacitor 1925 remains at least approximately constantat the desired output voltage.

It is also known to provide a multi-phase DC-to-DC buck converter 2000including a plurality of interleaving output phases 2005 a, 2005 b, 2005c, . . . , 2005 n, as shown in FIG. 20. As shown in FIG. 20, each of theoutput phases 2005 a, 2005 b, 2005 c, . . . , 2005 n is assigned arespective switching arrangement, including a high-side switch, alow-side switch, and an output inductor. In operation, the controlcircuit 2010 periodically operates the output phases 2005 a, 2005 b,2005 c, . . . . , 2005 n in a time-delayed sequence.

By operating the output phases 2005 a, 2005 b, 2005 c, . . . , 2005 n ina phase-delayed sequence, the conventional multi-phase buck converter2000 distributes current production across the multiple output phases2005 a, 2005 b, 2005 c, . . . , 2005 n, thereby distributing heatgeneration and reducing the requirements for the output capacitor 1925,such that a smaller output capacitor 125 may be utilized.

However, since conventional multi-phase buck converters require a fixednumber of point-to-point connections between the control circuit 2010and the output phases 2005 a, 2005 b, 2005 c, . . . , 2005 n,conventional multi-phase buck converters do not provide a robustarchitecture capable of easy expandability to include any number ofdesired phases.

Furthermore, conventional multi-phase buck converters do not optimallycontrol the output voltage in response to a request for a lower desiredoutput voltage or a decrease in current demand of the load 1935. By notoptimally controlling the output voltage, conventional multi-phase buckconverters may produce unwanted voltage spikes, which may damagecircuitry connected to the output of the buck converter.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a multi-phase buckconverter that overcomes the disadvantageous of prior art buckconverters described above. To achieve this object, the presentinvention provides a multi-phase buck converter for producing an outputvoltage to a load, the output voltage being produced from an inputvoltage in accordance with a desired voltage, the converter including anoutput capacitor, the output voltage being provided by the outputcapacitor; a plurality of output switch arrangements having respectiveoutput inductors coupled to the output capacitor, the switcharrangements being controllable to provide respective phase outputcurrents to the output capacitor through the respective outputinductors; a plurality of phase output arrangements respectively coupledto the output switch arrangements, the phase output arrangements beingcontrollable to set the respective phase output currents supplied by theoutput switch arrangements; a phase control bus communicatively coupledto each of the phase output arrangements; and a phase controlarrangement communicatively coupled to the phase control bus, the phasecontrol arrangement being configured to control the phase outputarrangements to set the respective phase output currents supplied by theoutput switch arrangements so that the output voltage approximates or isregulated to the desired voltage, in which the phase control arrangementand the phase output arrangements are provided as respective integratedcircuits, and the phase control arrangement is configured to control thephase output arrangements via the phase control bus.

By separating the functions of the phase control arrangement and thephase output arrangements, an exemplary multi-phase buck converteraccording to the present invention contains no unused or redundantsilicon, since the buck converter may include only those number of phaseoutput arrangements required for a particular application. Thus, if adesign engineer requires, for example, a three-phase buck converter fora particular application, the engineer may design the multi-phase buckconverter to include only three phase output arrangements, each of whichis assigned to a respective one of the three phase outputs. Furthermore,the phase control bus (e.g., a 5-wire analog bus) permits themulti-phase buck converter of the present invention to communicate witha potentially unlimited number of phase output arrangements, withoutrequiring point-to-point electrical connections between the phasecontrol arrangement and each of the phase output arrangements. In thismanner, the multi-phase buck converter permits an efficient and easilyscalable phase architecture.

In accordance with another exemplary embodiment of the presentinvention, the multi-phase buck converter is provided with a phase errordetect arrangement configured to produce a phase error signal if a phaseoutput arrangement is incapable of providing a phase output current tomatch the average inductor current of the phase output arrangements. Inthis manner, the phase control arrangement is provided with a signal fordetecting a defective phase and, if appropriate, may deactivate thedefective phase and/or enable a back-up phase output arrangement.

In accordance with yet another exemplary embodiment of the presentinvention, each of the output phase arrangements operates to switch offboth the high-side and low-side switches in response to a request for alower desired output voltage (V_(DES)) or a decrease in current demandof the load. In this manner, the slew rate of the inductor is increased,which enhances the response time of the multi-phase buck converter ofthe present invention and prevents disadvantageous negative currentsfrom flowing through the output inductor and possibly damaging the powersupply.

In accordance with still another exemplary embodiment of the presentinvention, each of the output phase arrangements includes a currentsense amplifier, a resistor R_(CS) electrically connected between thepositive input of the current sense amplifier and an output inductornode, and a capacitor C_(CS) electrically connected between the positiveand negative inputs of the current sense amplifier, with the outputinductor also being connected to the negative input of the current senseamplifier.

By connecting resistor R_(CS) and capacitor C_(CS) across the nodes ofthe output inductor, the current flowing through the output inductor 220may be sensed by selecting resistor R_(CS) and capacitor C_(CS) suchthat the time constant of resistor R_(CS) and capacitor C_(CS) equalsthe time constant of the output inductor 220 and its DC resistance(i.e., inductance L/inductor DCR, where DCR is the inductor DCresistance), the voltage across capacitor. In this manner, thisembodiment of the present invention permits each of the output phasearrangements to sense the current provided to the load in a losslessmanner (i.e., without interfering with the current provided to theload).

In accordance with yet another exemplary embodiment of the presentinvention, the phase control arrangement includes droop circuitryconfigured to reduce the output voltage in proportion to the currentdemand of the load. In this manner, this exemplary embodiment permits anefficient and simple method to adaptively modify the output voltage viaadaptive voltage positioning.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an exemplary buck converter according tothe present invention.

FIG. 2 is a block diagram of an output switch arrangement according tothe present invention.

FIG. 3 is a block diagram showing the phase control arrangement of FIG.1 in greater detail.

FIG. 4 is a graph showing the response of an exemplary buck converteraccording to the present invention in response to a load-step decrease.

FIG. 5 is a block diagram showing the phase timing arrangement of FIG. 3in greater detail.

FIG. 6 is a block diagram showing the PWM arrangement of FIG. 3 ingreater detail.

FIG. 7 is a block diagram showing a variant of the exemplary PWMarrangement of FIG. 6 configured to reduce the output voltageproportionally to an increase in load current.

FIG. 8 is a block diagram showing another exemplary phase controlarrangement according to the present invention.

FIG. 9 a is a graph showing an exemplary periodic charge cycle durationfor an output switch arrangement.

FIG. 9 b is a graph showing an output switch arrangement control inresponse to a request for a lower desired output voltage.

FIG. 10 is a block diagram of an exemplary phase output arrangementaccording to the present invention.

FIG. 11 is a block diagram of an exemplary start-time arrangementaccording to the present invention.

FIG. 12 a is a graph showing an exemplary phase timing signal accordingto the present invention.

FIG. 12 b is a graph showing the phase timing signal of FIG. 12 offsetby a set-point voltage value.

FIG. 12 c is a graph showing the output of a phase time comparator.

FIG. 12 d is a graph showing phase timing for eight phases with respectto a triangular phase timing signal.

FIG. 12 e is a block diagram showing another exemplary start-timearrangement according to the present invention.

FIG. 13 is a block diagram showing an exemplary charge-on durationarrangement according to the present invention.

FIG. 14 is a block diagram showing an exemplary ramp generator accordingto the present invention.

FIG. 15 is a block diagram showing an exemplary current sensearrangement according to the present invention.

FIG. 16 is a block diagram showing an exemplary phase output arrangementaccording to the present invention implemented as a separate integratedcircuit.

FIG. 17 is a block diagram showing the connectivity between a phasecontrol arrangement and a plurality of phase output arrangementsaccording to the present invention.

FIG. 18 is a block diagram showing an exemplary over-temperature detectcircuit according to the present invention.

FIG. 19 is a block diagram showing a single phase buck converteraccording to the prior art.

FIG. 20 is a block diagram showing a multi-phase buck converteraccording to the prior art.

DETAILED DESCRIPTION

Referring now to FIG. 1, there is seen a first exemplary multi-phasebuck converter 100 according to the present invention. Buck converter100 includes phase control arrangement 105 electrically andcommunicatively coupled to input bus 130, phase output arrangements 110a, 110 b, 110 c, . . . , 110 n electrically and communicatively coupledto the phase control arrangement 105 via a phase control bus 115 (e.g.,a 5-wire analog bus), output switch arrangements 120 a, 120 b, 120 c, .. . , 120 n electrically and communicatively coupled to an input voltage(V_(IN)) and phase output arrangements 110 a, 110 b, 110 c, . . . , 110n, an output capacitor 125 electrically coupled to the output switcharrangements 120 a, 120 b, 120 c, . . . , 120 n for producing an outputvoltage (V_(OUT)) , and a load 135 electrically connected between theoutput voltage (V_(OUT)) and ground.

The exemplary multi-phase buck converter 100 of FIG. 1 may be used, forexample, in applications requiring small sizes, design flexibility,various low voltage outputs, high currents and fast transient responses,and the buck converter 100 may include one or more output phases, forexample, three phases, each of which may be implemented by a respectiveone of the phase output arrangements 110 a, 110 b, 110 c, . . . , 110 n.

The control arrangement 105 includes circuitry configured to control thephase output arrangements 110 a, 110 b, 110 c, . . . , 110 n bycommunicating phase control signals via the phase control bus 115, sothat the phase output arrangements 110 a, 110 b, 110 c, . . . , 110 nproduce the output voltage (V_(OUT)) in accordance with a desired outputvoltage variable (V_(DES)), which may be provided to the controlarrangement 105 via input bus 130.

Each of the phase output arrangements 110 a, 110 b, 110 c, . . . , 110 nincludes circuitry configured to control respective output switcharrangements 120 a, 120 b, 120 c, . . . , 120 n in response to the phasecontrol signals communicated by the control arrangement 105 via thephase control bus 115. For this purpose, the phase output arrangements110 a, 110 b, 110 c, . . . , 110 n operate to control the respectiveswitch arrangements 120 a, 120 b, 120 c, . . . , 120 n to produce theoutput voltage (V_(OUT)) in accordance with the desired output voltagevariable (V_(DES)).

Referring now to FIG. 2, there is seen an exemplary output switcharrangement 120 n according to the present invention. Output switcharrangement 120 n includes a high-side switch 205 and a low-side switch210 (e.g., transistor switches, FET switches, FET rectifier, etc)electrically connected to one another via an inductor node 215. Theinput voltage (V_(IN)) is electrically connected to the high-side switch205 and a ground voltage is electrically connected to the low-sideswitch 210. The output voltage (V_(OUT)) is produced at an output-nodeside 220 a of an output inductor 220, which is also electricallyconnected to the switch node 215.

In operation, the high-side and low-side switches 205, 210 of switcharrangement 120 n are controlled by the phase output arrangement 110 nto produce the desired output voltage (V_(OUT)) at the output-node side220 a of the output inductor 220. For this purpose, the high-side switch205 is initially switched on, while the low-side switch 210 remains off.This causes a voltage drop across the output inductor 220 ofapproximately (V_(IN)−V_(OUT)), which causes a current to build insidethe output inductor 220. At a subsequent time, the high-side switch 205is switched off, and the low-side switch 210 is switched on. Since thecurrent within the inductor 220 cannot change instantly, the currentcontinues to flow through the output inductor 220, thereby charging theoutput capacitor 125 and causing the voltage drop across the outputcapacitor 125 to rise.

In this manner, the high-side and the low-side switches 205, 210 may besuitably switched controlled at appropriate times, until the voltagedrop across the output capacitor 125 equals the desired output voltage(V_(DES)). Once the desired output voltage (V_(DES)) is reached, thehigh-side and the low-side switches 205, 210 may be periodicallycontrolled so that the output inductor 220 provides an amount of currentequal to the current demand of the load 135 connected across the outputcapacitor 125. By providing no more and no less than the current demandof the load 135, the voltage drop (V_(OUT)) across the output capacitor125 remains approximately constant at the desired output voltage(V_(DES)).

In accordance with the exemplary embodiment of the present inventiondescribed above, the output phase arrangement 110 n controls thehigh-side and low-side switches 205, 210 during a periodic charge cycleduration, which may be characterized by an assigned phase delay, aperiodic start time, and a charge-on duration. Referring now to FIG. 9a, there is seen an exemplary periodic charge cycle duration 900 for theoutput switch arrangement 120 n, including an assigned phase delay 905,a periodic start time 910, and a charge-on duration 915. As shown inFIG. 9 a, the high-side switch 205 is switched on at the periodic starttime 910, remains on during the charge-on duration 915, and is switchedoff at the end of the charge-on duration 915. After the charge-onduration 915 expires, the high-side switch remains off for the remainderof the periodic charge cycle duration 900. During normal operation, thelow-side switch 210 is controlled such that the low-side switch 210 isswitched on when the high-side switch is switched off, and vice versa.In this manner, the output inductor 220 builds up current during thecharge-on duration 915 and releases at least a portion of the currentafter the charge-on duration 915 during the remainder of the periodiccharge cycle duration 900.

By controlling the high-side and low-side switches 205, 210 in themanner described above, the amount of current built up in the outputinductor 220 may be controlled by changing the charge-on duration 915relative to the periodic charge cycle duration 900. For example, if thecharge-on duration 915 is equal to half the periodic charge cycleduration 900 (i.e. , 50% duty cycle), the switch arrangement 120 n willprovide the output capacitor 125 with half the maximum current of thebuck converter 100. Or, for example, if the charge-on duration 915 isequal to the periodic charge cycle duration 900 (i.e., 100% duty cycle),the switch arrangement 120 n will provide the output capacitor 125 withthe maximum current of the buck converter 100.

During normal operation, the low-side switch 210 is controlled indichotomy with the high-side switch 205. That is, when the high-sideswitch 205 is switch on, the low-side switch 210 is switched off, andvice versa. In this manner, one of the high-side and low-side switches205, 210 is on at all times. However, in response to certain operatingconditions, it may be desirous to switch off both switches 205, 210.

Therefore, in accordance with another exemplary embodiment of thepresent invention, the output phase arrangement 110 n operates to switchoff both the high-side and low-side switches 205, 210 in response to theoccurrence of either of two unique operating conditions: a request for alower desired output voltage (V_(DES)) or a decrease in current demandof the load 135 drop (i.e., a load-step decrease).

A request for a lower desired output voltage (V_(DES)) may causenegative inductor currents to flow through the output inductor 220.Negative currents transform the buck converter 100 into a boostconverter by transferring energy from the output capacitor 125 to theinput voltage (V_(IN)). This energy may damage the power supply (notshown) and/or other components, may cause the voltage control loop tobecome unstable, and may result in wasted energy.

As shown in FIG. 9 b, to prevent the creation of negative inductorcurrents, both the high-side and low-side switches 205, 210 are turnedoff in response to a request for a lower desired output voltage(V_(DES)) . In this manner, the current built up in the output inductor220 is discharged through the load 135, rather than through the powersupply.

As the current discharges through the load 135, the output voltage(V_(OUT)) across the output capacitor 125 drops. Once the output voltage(V_(OUT)) drops to approximately the lower desired output voltage(V_(DES)), negative currents are no longer a concern, and the high-sideand low-side switches 205, 210 may be operated in normal fashion.

When the current demands of the load 135 drop (i.e., a load-stepdecrease), the high-side and low-side switches 205, 210 should becontrolled to reduce the current supplied to the output capacitor 125 bythe output inductor 220. However, in conventional buck converters, theminimum time required to reduce the current (i.e. , a current transient)in the output inductor 220 in response to a load-step decrease isgoverned by the following equation:T _(SLEW) =[L×(I _(MAX) −I _(MIN))]/V _(OUT),   (1)where the high-side and low-side switches 205, 210 are implemented asFET rectifiers.

Thus, when the current demands of the load decrease, the currenttransient (i.e. , the current built up inside the output inductor at thetime of a load-step decrease) of the output inductor 220 of conventionalbuck converters will cause the output capacitor 125 voltage to rise.Although the current demand of the load 135 will eventually drain theexcess charge of the output capacitor 125, the short-time durationvoltage-spike on the output voltage (V_(OUT)) may damage sensitivecircuitry connected to the buck converter 100.

However, in accordance with an exemplary embodiment of the presentinvention, the output phase arrangement 110 n operates to switch offboth the high-side and low-side switches 205, 210 (i.e. , body-brake) inresponse to a decrease in current demand of the load 135 drop (i.e. , aload-step decrease). In this manner, the slew rate (i.e. , the rate atwhich current may be reduced) of the output inductor 220 may besignificantly increased, where the high-side and low-side switches 205,210 are implemented as FET rectifiers. By turning off both the high-sideand low-side switches 205, 210, the switch node voltage is forced todecrease until the body diode of the FET rectifier conducts. Thisincreases the voltage across the inductor from V_(OUT) to V_(OUT)+ thevoltage across the body diode (i.e. , V_(BODY DIODE)). Thus, the slewrate of the output inductor 220 is reduced in accordance with thefollowing equation:T _(SLEW) =[L×(I _(MAX) −I _(MIN))](V _(OUT) +V _(BODY DIODE))   (2)

Therefore, in accordance with this exemplary embodiment of the presentinvention, the current transient built up inside the output inductor 220during a load-step decrease condition may be drained off more rapidly,thereby causing a much less pronounced voltage spike, when compared tothe prior art, as shown in FIG. 4. In fact, since the voltage dropacross the body diode may be higher than the output voltage V_(OUT), theinductor current slew rate may be increased by two times or more.

Referring now to FIG. 10, there is seen an exemplary phase outputarrangement 110 n according to the present invention for controlling thehigh-side and low-side switches 105, 110 of the output switcharrangement 120 n in the manner described above. Phase outputarrangement 110 n includes a start-time arrangement 1005, a charge-onduration arrangement 1010, a current sense arrangement 1015 electricallycoupled to the charge-on duration arrangement 1010, an S-R latch 1020electrically coupled to the start-time arrangement 1005 and thecharge-on duration arrangement 1010, and an and-gate 1025 electricallycoupled to the S-R latch 1020 and the charge-on duration arrangement1010.

The start-time arrangement 1005 includes circuitry configured todetermine the periodic start time 910 and the phase delay 905 shown inFIG. 9 a. For this purpose, the start-time arrangement 1005 receives aphase timing signal 1030 from the phase control arrangement 105. Thephase timing signal 1030 may include, for example, a periodic analogsignal having a period equal to the periodic charge cycle duration 900(e.g. , a periodic saw-tooth waveform, a periodic sinusoidal waveform, aperiodic triangular waveform, etc). Using the periodic analog signal1030, the start-time arrangement 1005 may determine the periodic starttime 910 and the phase delay 905, and generate a periodic clock pulse1035 at the periodic start time 910. The clock pulse 1035 sets the S-Rlatch 1020, causing the high-side switch 205 to switch on and thelow-side switch 210 to switch off at the beginning of the charge-onduration 915.

The charge-on duration arrangement 1010 includes circuitry configured todetermine the charge-on duration 915, to reset the S-R latch 1020 at theend of the charge-on duration 915, and to switch of both the high-sideand low-side switches 205, 210 in response to a request for a lowerdesired output voltage (V_(DES)) or a decrease in current demand of theload 135 drop (i.e. , a load-step decrease). For this purpose, thecharge-on duration arrangement 1010 receives a Pulse-Width-Modulation(PWM) control signal 1040 from the phase control arrangement 105. ThePWM control signal 1040 may include, for example, an analog signalhaving a value proportional to the difference between the desired outputvoltage (V_(DES)) and the actual output voltage (V_(OUT)). Using the PWMcontrol signal 1040, the charge-on duration arrangement 1010appropriately determines the charge-on duration 915 for the high-sideand low-side switches 205, 210. Furthermore, the charge-on durationarrangement 1010 is configured to modify the charge-on duration 915 inaccordance with the amount of current supplied to the output capacitor125 by the output inductor 220. For this purpose, the charge-on durationarrangement 1010 receives a current difference signal 1050 from thecurrent-sense arrangement 1015 characterizing the amount of currentsupplied by the output inductor 220 relative to the average current 1045provided by all the output switch arrangements 120 a, 120 b, 120 c, . .. , 120 n, so that the charge-on duration arrangement 1010 may increasethe charge-on duration 915 if the amount of current supplied by theoutput inductor 220 is less than the average current 1045 provided byall the output switch arrangements 120 a, 120 b, 120 c, . . . , 120 n.By increasing the charge-on duration 915, the output inductor 220supplies more current to the output capacitor 125. After the charge-onduration 915 expires, the charge-on duration arrangement 1010 resets theS-R latch 1020, which causes the high-side switch 205 to switch off andthe low-side switch 210 to switch on for the remainder of the periodiccharge cycle duration 900.

In response to a request for a lower desired output voltage (V_(DES)) ora decrease in current demand of the load 135 drop (i.e. , a load-stepdecrease), which may be determined from the PWM control signal 1040communicated by the phase control arrangement 105, the charge-onduration arrangement 1010 operates to turn off both the high-side andlow-side switches 205, 210. For this purpose, the charge-on durationarrangement 1010 resets the S-R latch 1020 and transmits a logical “0”to the and-gate 1025, thereby causing both the high-side and low-sideswitches 205, 210 to switch off.

The S-R latch 1020 is reset dominant allowing all phase outputarrangements 110 a, 110 b, 110 c, . . . , 110 n to go to zero duty cyclewithin a few tens of nanoseconds. Phases may overlap and go to 100% dutycycle in response to a load step increase with the turn-on gated byclock pulses. In this manner, this method of controlling the phaseoutput arrangements 110 a, 110 b, 110 c. . . , 110 n provides a “singlecycle transient response,” in which the output inductor 220 currentchanges in response to load transients within a single switching cycle,thereby maximizing the effectiveness of the power train and minimizingthe requirements of the output capacitor 125.

The current sense arrangement 1015 includes circuitry configured togenerate the current difference signal 1050 for modifying the charge-onduration 915 in accordance with the current flowing through the outputinductor 220 relative to the average current 1045 provided by all theoutput switch arrangements 120 a, 120 b, 120 c, . . . , 120 n.

Referring now to FIG. 11, there is seen an exemplary start-timearrangement 1005 according to the present invention for generating theclock pulse 1035 in accordance with the periodic start time 910 and thephase delay 905. Start-time arrangement 1005 includes a phase timingcomparator 1105 and a one-shot pulse generator 1110 electricallyconnected to the output of the phase timing comparator 1105. In thisexemplary embodiment, the phase timing signal 1030 is a periodictriangular waveform 1030 having a period equal to the periodic chargecycle duration 900 and an amplitude varying between 0 volts and 5 volts,as shown in FIG. 12 a.

Referring now to FIG. 12 b, there is seen a timing diagram showing theoutputs of the phase timing comparator 1105 and the one-shot pulsegenerator 1110. As shown in FIG. 12 b, the output of the phase timingcomparator 1105 is equal to the phase timing signal 1030 offset by aconstant set-point voltage 1115. Thus, the output of the phase timingcomparator 1105 crosses the zero-voltage axis once in the positivedirection during the periodic charge cycle duration 900 at a time equalto the phase delay 905, thereby causing the one-shot pulse generator1110 to generate the clock pulse 1035.

By appropriately selecting the set-point voltage 1115 between 0 and 5volts, the one-shot pulse generator 1110 may be controlled to generatethe clock pulse 1035 at any time during the first half 900 a of theperiodic phase cycle duration 900. To cause the one-shot pulse generator1110 to generate the clock pulse 1035 during the second half 900 b ofthe periodic phase cycle duration 900, the inputs to the phase timingcomparator 1105 may be switched, such that the phase timing signal isprovided to the negative input of the phase timing comparator 1105 andthe set-point voltage 1115 is provided to the positive input of thephase timing comparator 1105. In this manner, the outputs of the phasetiming comparator 1105 and the one-shot pulse generator 1110 resemblethose shown in the timing diagram of FIG. 12 c.

Thus, in accordance with the present invention, each of the phase outputarrangements 110 a, 110 b, 110 c, . . . , 110 n may be assigned a uniquephase delay 905 and periodic start time 910 during the periodic phasecycle duration 900, without requiring separate point-to-point electricalconnections between the phase control arrangement 105 and the phaseoutput arrangements 110 a, 110 b, 110 c, . . . , 110 n. Furthermore, ifthe phase output arrangements 110 a, 110 b, 110 c, . . . , 110 n are tobe implemented using separate phase integrated circuits, an especiallyefficient and simple assignment of the phase delay 905 and periodicstart time 910 for each of the phase output arrangements 110 a, 110 b,110 c, . . . , 110 n may be effected if both inputs of the phase timingcomparator 1105 are electrically connected to input pins of a respectivephase integrated circuit.

Referring now to FIG. 12 d, there is seen a time diagram showing theoutputs of respective one-shot pulse generators for an exemplary buckconverter 100 according to the present invention having eight phaseoutput arrangements 110 a, 110 b, 110 c, . . . , 110 h.

Referring now to FIG. 12 e, there is seen the start-time arrangement1005 of an exemplary phase output arrangement 110 n implemented as aseparate and distinct phase IC 1250. As shown in FIG. 12 e, the phase ICincludes electrical contact pins 1255 a and 1255 b electricallyconnected to the inputs of the phase timing comparator 1105,respectively. A voltage divider is provided between a reference voltage1270 and ground, the voltage divider comprising resistors 1265 a and1265 b connected to one another at node 1260. By suitably selectingresistors 1265 a and 1265 b, a predetermined set-point voltage 1115 maybe provided to the phase timing comparator 1105 via electrical contactpin 1255 b.

Referring now to FIG. 13, there is seen an exemplary charge-on durationarrangement 1010 according to the present invention. Charge-on durationarrangement 1010 includes charge-on duration amplifier 1305, body-brakedetect amplifier 1315, a fractional multiplier 1320 electricallyconnected to the negative input of the body-brake detect amplifier 1315,and a ramp generator 1310 electrically coupled to the negative input ofthe charge-on duration amplifier 1305 and to the fractional multiplier1320.

At a time before the start-time arrangement 1005 produces clock pulse1035 to set the S-R latch 1020, the inverted output 1020 a of the S-Rlatch 1020 asserts a logical high level “1” on the reset line of theramp generator 1310 of the charge-on duration arrangement 1010. Thiscauses the ramp generator 1310 to generate a constant default outputvoltage on ramp output line 1325 (the constant default voltage is alsopermanently provided on default voltage output line 1330). After thestart-time arrangement 1005 sets the S-R latch 1020, the high-sideswitch 205 is switched on and the inverted output 1020 a of the S-Rlatch 1020 asserts a logical low level “0” on the reset line of the rampgenerator 1310, causing the voltage on the ramp output line 1325 to rampup from the default output voltage. The charge-on duration amplifier1305 compares the ramp output line 1325 to the PWM control signal 1040,which, in this exemplary embodiment of the present invention, is ananalog voltage signal proportional to the difference between the desiredoutput voltage (V_(DES)) and the actual output voltage (V_(OUT))(V_(DES)−V_(OUT)) . Once the voltage at the ramp output line 1325reaches the PWM control signal 1040 voltage level, the charge-onduration amplifier 1305 causes the S-R latch 1020 to reset, which causesthe high-side switch 205 to switch off and causes the inverted output1020 a of the S-R latch 1020 to assert a logical high level “1” on thereset line of the ramp generator 1310 to reset the ramp output line 1325to the default voltage.

In this manner, the charge-on duration 915 represents the time betweenwhen the start-time arrangement 1005 produces the clock pulse 1035 andwhen the ramp output line 1325 of the ramp generator 1310 equals the PWMcontrol signal 1040 voltage level. Thus, the greater the deviationbetween the actual output voltage (V_(OUT)) and the desired outputvoltage (V_(DES)), the greater the PWM control signal 1040 voltagelevel, and thus the greater the charge-on duration 915.

Furthermore, the charge-on duration arrangement 1010 may modify thecharge-on duration 915 in accordance with the amount of current suppliedto the output capacitor 125 by the output inductor 220. For thispurpose, the ramp generator 1310 receives a current difference signal1050 from the current-sense arrangement 1015 that characterizes theamount of current supplied by the output inductor 220 relative to theaverage current 1045 provided by all the output switch arrangements 120a, 120 b, 120 c, . . . , 120 n. For example, the current differencesignal 1050 may provide a voltage value in proportion to the differencebetween the current supplied by the output inductor and the averagecurrent supplied by all the output switch arrangements 120 a, 120 b, 120c, . . . , 120 n. Using the current difference signal 1050, the rampgenerator 1310 may vary the rate at which the voltage at the output line1325 ramps up, so that the rate at which the voltage at the ramp outputline 1325 ramps up decreases as the difference between the currentsupplied by the output inductor and the average current supplied by allthe output switch arrangements 120 a, 120 b, 120 c, . . . , 120 nincreases.

Thus, if the amount of current supplied by the output inductor 220 isless than the average current 1045 provided by all the output switcharrangements 120 a, 120 b, 120 c, . . . , 120 n, the reduced ramp-uprate of the voltage at the ramp output line 1325 will cause thecharge-on duration 915 to increase, thereby causing the output inductor220 to supply more current to the output capacitor 125.

The charge-on duration arrangement 1010 is also configured to switch offboth the high-side and low-side switches 205, 210 in response to arequest for a lower desired output voltage (V_(DES)) or a decrease incurrent demand of the load 135 drop (i.e. , a load-step decrease). Forthis purpose, the fractional multiplier 1320 produces a fractionalmultiple (e.g. , 90%) of the default voltage of the ramp generator 1310,and provides the fractional multiple to the body-brake detect amplifier1315. The body-brake detect amplifier 1315 compares the fractionalmultiple of the default voltage with the PWM control signal 1040 voltagelevel (i.e. , a voltage level in proportion to V_(DES)−V_(OUT)) andgenerates a signal to switch off the high-side and low-side switches205, 210 if the PWM control signal 1040 voltage level drops below thefractional multiple of the default voltage.

It should be appreciated that various conditions may cause thebody-brake detect amplifier 1315 to switch off the high-side andlow-side switches 205, 210. For example, a sudden decrease in currentdemand of the load 135, which would cause V_(OUT) to rise in relation toV_(DES), may cause the PWM control signal 1040 voltage level to dropbelow the fractional multiple of the default voltage. Alternatively, forexample, the phase control arrangement 105 may force the PWM controlsignal 1040 below the fractional multiple of the default voltage inresponse to a request for a decrease in the desired output voltage(V_(DES)), as more fully described below.

Referring now to FIG. 14, there is seen an exemplary ramp generator 1310according to the present invention. Ramp generator 1310 includes a clampcircuit 1405 and a programmable current source 1410 electricallyconnected to the ramp output line 1325. The clamp circuit 1405 includesan operational amplifier 1415 and a clamp diode 1420, both of whichoperate together to force the ramp output line 1325 to the defaultvoltage when the enable input 1415 a of the operational amplifier 1415is asserted.

Ramp generator 1310 also includes an phase error detect amplifier 1450,a fractional multiplier 1455 electrically connected to the phase errordetect amplifier 1450 and the default voltage, and a switch 1460electrically connected to the output of the phase error detect amplifier1450, all of which work together to generate a phase error signal 1465if the phase output arrangement 120 n is not capable of providing enoughcurrent to match the average current 1045 provided by the output switcharrangements 120 a, 120 b, 120 c, . . . , 120 n. Using the phase errorsignal 1465, the buck converter 100 may deactivate the damaged phaseoutput arrangement 120 n and/or activate a backup phase outputarrangement 120 n.

At a time before the start-time arrangement 1005 produces clock pulse1035 to set the S-R latch 1020, the inverted output 1020 a of the S-Rlatch 1020 asserts a logical high level “1” on the reset line of theramp generator 1310, which enables the clamp circuit 1405, therebyclamping the voltage at the ramp output line 1325 to the defaultvoltage. After the start-time arrangement 1005 sets the S-R latch 1020,the high-side switch 205 is switched on and the inverted output 1020 aof the S-R latch 1020 asserts a logical low level “0” on the reset lineof the ramp generator 1310, which disables the clamp circuit 1405. Withthe clamp circuit 1405 disabled, the ramp capacitor 1425 receivescurrent from V_(IN) through the ramp resistor 1430, thereby causing thevoltage at the ramp output line 1325 of the ramp generator 1310 to rampup. Once the voltage at the output line 1325 reaches the PWM controlsignal 1040 voltage level, the charge-on duration amplifier 1305 causesthe S-R latch 1020 to reset, which causes the high-side switch 205 toswitch off and the inverted output 1020 a of the S-R latch 1020 toassert a logical high level “1” on the reset line of the ramp generator1310, thereby causing the clamp circuit 1405 to clamp the output line1325 to the default voltage.

The ramp-up time of the voltage on the ramp output line 1325 of the rampgenerator 1310 may be modified in accordance with the amount of currentthe output inductor 220 supplies to the output capacitor 125 bycontrolling the programmable current source 1410 with the currentdifference signal 1050 generated by the current sense arrangement 1015.For this purpose, the current source 1410 may be controlled to sink anamount of current from the ramp output line 1325 proportional to thedifference between the current supplied by the output inductor 220 andthe average current supplied by all the output switch arrangements 120a, 120 b, 120 c, . . . , 120 n. By removing (i.e. , sinking) currentfrom the ramp output line 1325, the ramp capacitor 1425 charges moreslowly, thereby causing the voltage at the ramp output line 1325 to rampup at a slower rate.

By charging the ramp capacitor 1425 from V_(IN) through the rampresistor 1430, the ramp-up rate of the voltage at the ramp output line1325 will automatically compensate for changes in the input voltageV_(IN) I which may occur, for example, due to variations in the outputvoltage of the power supply (not shown) or due to voltage drops in theprinted circuit board (PCB) related to changes in load current.

Furthermore, in accordance with another exemplary embodiment of thepresent invention, the desired output voltage (V_(DES)) is used as thedefault voltage of the ramp generator 1310. Since the desired outputvoltage (V_(DES)) is a relatively stable voltage level produced from aD/A converter inside the phase control arrangement 105, the desiredoutput voltage (V_(DES)) does not fluctuate between different phaseoutput arrangements 110 a, 110 b, 110 c, . . . , 110 n. In this manner,differences in ground or input voltages at the phase output arrangements110 a, 110 b, 110 c, . . . , 110 n have little or no effect on the rampvoltage output of the ramp generator 1310, since the voltage of theoutput line 1325 is referenced to the desired output voltage (V_(DES)).

If the phase output arrangement 120 n is damaged or otherwiseinoperative, the current supplied by the output inductor 220 may drop toa level at which the current source 1410 sinks current at a faster ratethan the ramp capacitor 1425 charges. In this case, the ramp outputsignal 1325 may begin to ramp downwards in voltage, causing the phaseerror detect amplifier 1450 to trigger the switch 1460 and produce aphase error signal, which may be used to deactivate the damaged phaseoutput arrangement 120 n and/or activate a backup phase outputarrangement 120 n.

Referring now to FIG. 15, there is seen an exemplary current sensearrangement 1015 according to the present invention. The current sensearrangement 1015 includes circuitry configured to generate a currentdifference signal 1050 characterizing the difference between the currentsupplied by the output inductor 220 and the average current supplied byall the output switch arrangements 120 a, 120 b, 120 c, . . . , 120 n.For this purpose, he current sense arrangement 1015 includes an inductorcurrent detection arrangement 1505 configured to produce an inductorcurrent signal 1510 in proportion to the amount of current flowingthrough the output inductor 220. The inductor current detectionarrangement 1505 includes a current sense amplifier 1515, a resistorR_(CS) electrically connected between the positive input of the currentsense amplifier 1515 and output inductor node 215, and a capacitorC_(CS) electrically connected between the positive and negative inputsof the current sense amplifier 1515, with inductor node 220 a also beingconnected to the negative input of the current sense amplifier 1515.

By connecting resistor R_(CS) and capacitor C_(CS) across the nodes 215,220 a of the output inductor 220, the current flowing through the outputinductor 220 may be sensed in accordance with the following equation:

$\begin{matrix}{{V_{C}(s)} = {{{V_{L}(s)}\frac{1}{1 + {{sR}_{CS}C_{CS}}}} = {{i_{L}(s)}\frac{R_{L} + {sL}}{1 + {{sR}_{CS}C_{CS}}}}}} & (3)\end{matrix}$By selecting resistor R_(CS) and capacitor C_(CS) such that the timeconstant of resistor R_(CS) and capacitor C_(CS) equals the timeconstant of the output inductor 220 (i.e. , inductance L/inductor DCR) ,the voltage across capacitor C_(CS) is proportional to the currentthrough the output inductor 220, and the inductor current detectionarrangement 1505 may be treated as if only a sense resistor with a valueof RL was used. A mismatch of time constants does not affect themeasurement of the inductor DC current, but does affect the AC componentof the current flowing through the output inductor 220.

Sensing the current flowing through the output inductor 220 may beadvantageous with respect to high-side and/or low-side sensing, sincethe actual output current delivered to the load 135 may be obtainedrather than a peak or sampled value of switch currents. Thus, the outputvoltage (V_(OUT)) may be positioned to meet a load line based on realtime information. In this manner, a current sense circuit according tothe present invention may advantageously support a single cycletransient response.

The current sense amplifier 1515 may be designed with a variable gainthat decreases with decreasing temperature, and a nominal gain, forexample, of 35 at 25 degrees Celsius and 31 at 125 degrees Celsius. Thiscorrelation of gain with temperature may compensate for a ppm/DegreesCelsius increase in the DCR of the output inductor 220.

The current sense amplifier 1515 communicates the current differencesignal 1510 to a current comparator 1520, which compares the currentsignal 1510 to the average inductor current 1045 of all phases toproduce the current difference signal 1050 for communication to thecharge-on duration arrangement 1010.

Current average resistor 1525 is provided between the current signal1510 and the average inductor current signal 1045. Since each of thephase output arrangements 110 a, 110 b, 110 c, . . . , 110 n provides asimilar current average resistor between their respective currentsignals and the average inductor current signal 1045, the averageinductor current signal 1045 exhibits a voltage in proportion to theaverage of the respective current signals of the phase outputarrangements 110 a, 110 b, 110 c, . . . , 110 n.

Referring now to FIG. 16, there is seen an exemplary phase outputarrangement 110 n and output switch arrangement 120 n according to thepresent invention. As shown in FIG. 16, like components are labeled withthe same reference characters as used in FIGS. 10 to 15. Additionally,the exemplary phase output arrangement 110 n of FIG. 16 provides asummation arrangement 1605 for adding the desired output voltage(V_(DES)) level to the sensed current signal, so that the default rampvoltage may be set to the desired output voltage (V_(DES)) level.

Referring now to FIG. 3, there is seen the exemplary multi-phase buckconverter 100 of FIG. 1, in which the phase control arrangement 105includes a phase timing arrangement 305 and a Pulse Width Modulation(PWM) arrangement 310 for generating the phase timing signal 1030 andthe PWM control signal 1040, respectively, via phase control bus 115(e.g. , a 5-wire analog bus). Phase control arrangement 105 alsoincludes additional circuitry arrangement 325 for generating additionalcontrol signals 330, which are not necessary for an understanding of thepresent invention.

Phase timing signal 1030 contains information to permit each of thephase output arrangements 110 a, 110 b, 110 c, . . . , 110 n todetermine its respective periodic start time 910, at which it mayoperate its respective one of the switch arrangements 120 a, 120 b, 120c, . . . , 120 n to provide current to the load 135. According to oneexemplary embodiment of the present invention, the phase timing signal1030 consists of a periodic voltage waveform, which is then decoded bythe phase output arrangements 110 a, 110 b, 110 c, . . . , 110 n, in amanner more fully described above.

Referring now to FIG. 5, there is seen an exemplary phase timingarrangement 305 according to the present invention for generating theperiodic phase timing signal 1030. Phase timing arrangement 305 includesa programmable oscillator arrangement 505 electrically coupled to aperiodic waveform generator 510, for example, a periodic triangularwaveform generator 510. The periodic triangular waveform generator 510is configured to generate the phase timing signal 1030 in accordancewith the frequency of the programmable oscillator arrangement 505, whichmay be varied by a frequency select input 515 of the input bus 130 or,alternatively, may be programmed by an external frequency selectresistor (not shown). In this manner, the frequency of the programmableoscillator arrangement 505 and, thus, the frequency of the periodicphase timing signal 1030, may be set to any desired frequency, forexample, a frequency in the range of 100 KHz to 1 MHz.

Referring back to FIG. 3, the PWM arrangement 310 of the phase controlarrangement 105 is configured to generate the PWM control signal 1040containing information and/or data to permit the phase outputarrangements 110 a, 110 b, 110 c, . . . , 110 n to determine a switch-onduration 915 for the high-side switch 205 of a respective one of theswitch arrangements 120 a, 120 b, 120 c, . . . , 120 n. As describedabove, the longer the switch-on duration 915 for the high-side switch205, the more current flows through the output inductor 220 of therespective switch arrangement. In this manner, the switch on duration915 may be dynamically controlled to compensate for changes in loadcurrent, transient load conditions, and/or a change in the desiredoutput voltage variable (V_(DES)).

Referring now to FIG. 6, there is seen an exemplary PWM arrangement 310according to the present invention for generating the PWM control signal1040. As shown in FIG. 6, PWM arrangement 310 includes adigital-to-analog converter (DAC) 605 configured to produce the desiredoutput voltage variable (V_(DES)) 610 from digital inputs 615 of theinput bus 130. High-gain error amplifier 620 compares the desired outputvoltage variable (V_(DES)) 610 with the actual output voltage (V_(OUT)),and generates an error signal 625 proportional to the difference betweenthe desired output voltage variable (V_(DES)) 610 and the actual outputvoltage (V_(OUT)). The error signal 625 may be communicated to the phasecontrol bus 115 as the PWM control signal 1040.

Since the PWM arrangement 310 of FIG. 6 generates a PWM control signal320 in proportion to the difference between the desired output voltagevariable (V_(DES)) 610 and the actual output voltage (V_(OUT)), the PWMcontrol signal 320 may be used by the phase output arrangements 110 a,110 b, 110 c, . . . , 110 n to keep the actual output voltage (V_(OUT))at the desired output voltage (V_(DES)) . In this manner, the PWMarrangement 310 and the phase output arrangements 110 a, 110 b, 110 c, .. . , 110 n form a closed loop for controlling the actual output voltage(V_(OUT)) irrespective of changes in load current.

For example, if the actual output voltage (V_(OUT)) drops below thedesired output voltage (V_(DES)) in response to an increase in loadcurrent, the switch-on duration 915 of the high-side switch 205 of arespective switch arrangement may be increased proportionally to the PWMcontrol signal 1040, thereby causing the output inductor 220 of therespective switch arrangement to supply more current to the outputcapacitor 125, which, in turn, causes the output voltage (V_(OUT)) torise. Alternatively, if the actual output voltage (V_(OUT)) rises abovethe desired output voltage (V_(DES)) in response to a decrease in loadcurrent, the switch-on duration of the high-side switch 205 of arespective switch arrangement may be decreased proportionally to the PWMcontrol signal 320, thereby causing the output inductor 220 of therespective switch arrangement to supply less current to the outputcapacitor 125, which, in turn, causes the output voltage (V_(OUT)) todrop.

The digital inputs 615 of the DAC 605 may include, for example, aplurality of Voltage-Identification (VID) digital signals generated byan external circuit, for example, a mobile Intel Pentium IVmicroprocessor. Voltage-Identification (VID) signals may be generated bythe microprocessor to communicate the voltage at which the processorcore should operate. In this manner, the digital-to-analog converter(DAC) 605 of the PWM arrangement 310 may generate the desired outputvoltage variable (V_(DES)) in accordance with the proper processor corevoltage.

Under certain circumstances, a request for a new desired output voltage(V_(DES)) may cause the digital inputs 615 (e.g. , the VID inputs) tochange during normal operation of the buck converter 100. When the phasecontrol arrangement 105 detects a change in the Voltage-Identification(VID) code, the phase control arrangement 105 may, for example, blankthe signals for a time duration, for example, 400 ns, to ensure that thedetected change is not due to skew or noise.

In response to a request for a higher desired output voltage (V_(DES)),the high-gain error amplifier 620 (via the PWM control signal 1040)causes the charge-on duration of the phase output arrangements 110 a,110 b, 110 c, . . . , 110 n to increase. Alternatively, in response to arequest for a lower desired output voltage (V_(DES)), the high-gainerror amplifier 620 causes the charge-on duration of the phase outputarrangements 110 a, 110 b, 110 c, . . . , 110 n to decrease. However, asdescribed above, a request for a lower desired output voltage (V_(DES))may cause disadvantageous negative currents to flow through the outputinductor 220.

Therefore, in accordance with another exemplary embodiment of thepresent invention, the phase control arrangement 105 is configured toswitch off the high-side and low-side switches 205, 210 of each of theoutput switch arrangements 120 a, 120 b, 120 c, . . . , 120 n inresponse to a request for a lower desired output voltage (V_(DES)). Forthis purpose, the PWM arrangement 310 may be provided with a step-downdetect arrangement 850, as shown in FIG. 8.

The step-down detect arrangement 850 detects a VID step-down conditionto prevent the negative inductor currents described above (i.e. , thenegative inductor currents associated with a request for a lower desiredvoltage). For this purpose, PWM arrangement 310 includes a clampingcircuit arrangement 855 configured to clamp the output of the high-gainerror amplifier 820 to a voltage level lower than the default voltage ofthe ramp generator 1310 of each of the phase output arrangements 110 a,110 b, 110 c, . . . , 110 n. In this manner, the PWM control signal 1040generated by the PWM arrangement 310 causes the charge-on durationarrangement 1010 of each of the phase output arrangements 110 a, 110 b,110 c, . . . , 110 n to switch off the high-side and low-side switches205, 210 of the respective output switch arrangements 120 a, 120 b, 120c, . . . , 120 n, until the output voltage (V_(OUT)) drops toapproximately the lower output voltage (V_(DES)).

In certain circumstances, adaptive voltage positioning may be requiredto reduce output voltage deviations during load transients and the powerdissipation when the load 135 is drawing maximum current. For thispurpose, the PWM arrangement 310 may include droop circuitry configuredto reduce the actual output voltage (V_(OUT)) proportionally to anincrease in load current.

Referring now to FIG. 7, there is seen a variant of the exemplary PWMarrangement 310 of FIG. 6 configured to reduce the output voltage(V_(OUT)) proportionally to an increase in load current. As shown inFIG. 7, the exemplary PWM arrangement 310 further includes droopcircuitry 700, which includes a current signal buffer electricallyconnected to the average inductor current signal 1045. In this exemplaryembodiment of the present invention, the average inductor current signal1045 is referenced to the desired output voltage variable (V_(DES)), sothat the output of the current signal buffer 705 is equal to(V_(DES)+I_(AVG)), where I_(AVG) is proportional to the average currentprovided by the output inductors 220 of the output switch arrangements120 a, 120 b, 120 c, . . . , 120 n. A droop resistor R_(VDRP) isprovided between the output of the current signal buffer 705 and thenegative input of the high-gain error amplifier 620, and an offsetresistor RFB is provided between the actual output voltage (V_(OUT)) andthe negative input of the high-gain error amplifier 620.

Thus, the voltage (v) at the negative input of the high-gain erroramplifier 620 is given by the following equation:

$\begin{matrix}{v = {{( {V_{DES} + I_{AVG}} )\frac{R_{FB}}{R_{FB} + R_{VDRP}}} + {V_{OUT}\frac{R_{VDRP}}{R_{FB} + R_{VDRP}}}}} & (4)\end{matrix}$

However, since the high-gain error amplifier 620 controls the voltageloop to keep its positive and negative inputs equal, the high-gain erroramplifier 620 operates to keep the voltage at its negative input equalto the desired output voltage (V_(DES)). Thus, the actual voltage(V_(OUT)) can be determined from the following equation:

$\begin{matrix}{V_{OUT} = {V_{DES} - {I_{AVG}\frac{R_{FB}}{R_{VDRP}}}}} & (5)\end{matrix}$

Thus, the exemplary PWM arrangement 310 of FIG. 6 operates to reduce theactual output voltage (V_(OUT)) proportionally to the average currentprovided by the output inductors 220 of the output switch arrangements120 a, 120 b, 120 c, . . . , 120 n. The positioning voltage (v) may beprogrammed by selecting an appropriate droop resistor R_(VDRP), so thatthe droop impendence produces the desired converter output impendence.

Referring now to FIG. 17, there is seen the exemplary buck converter 100implemented using discrete control and phase ICs. The exemplary buckconverter 100 of FIG. 17 includes a control IC 1705 containing all thefunctions of the phase control arrangement 105 and two phase ICs 1250 a,1250 b (see FIG. 16) containing all functions of the phase outputarrangements 110 a, 110 b, respectively.

Each of the control and phase ICs 1705, 1250 a, 1250 b may include anover-temperature detect circuit 1805, as shown in FIG. 18.Over-temperature detect circuit 1805 includes a VRHOT comparator 1810, aswitch 1815 electrically connected to the output of the VRHOT comparator1810, and a temperature sensing arrangement 1820 configured to produce avoltage proportional to the die temperature. Using an external pin 1825,the temperature threshold may be set using, for example, a voltagedivider connected to V_(IN). If the temperature of the die rises abovethe temperature threshold, the VRHOT comparator 1810 switches on theswitch 1815, thereby causing a VRHOT signal 1830 to be generated. TheVRHOT signal may be used, for example, to deactivate the phase or enableadditional phases to share in the current production burden.

1. A multi-phase switched voltage converter for providing an outputvoltage to a load, the output voltage being produced from an inputvoltage in accordance with a desired voltage, the converter comprising:an output circuit adapted to be coupled to the load; a plurality ofoutput switch units having respective output terminals coupled to theoutput circuit; the output switch units being controllable to providerespective phase output currents to the output circuit through therespective output terminals; a plurality of switch unit drivers in theform of separate integrated circuits respectively coupled to control theoutput switch units, the switch unit drivers being controllable todetermine the respective phase output currents supplied by the outputswitch units such that each switch unit driver controls a duty cycle ofswitches in the respective output switch unit to which it is coupled;and a master control unit in the form of a separate integrated circuitcoupled to drive each of the output switch unit drivers.
 2. The voltageconverter according to claim 1, further including a common control buswhich couples the master control unit to operate the output switch unitdrivers.
 3. The voltage converter according to claim 1, wherein each ofthe output switch units includes a high-side switch and a low-sideswitch coupled to the high-side switch at a respective switch node. 4.The voltage converter according to claim 3, wherein: each output switchunit includes an output inductor which couples the switch node of thatswitch unit to the output terminal; and the output circuit includes acapacitor coupled to each of the output terminals.
 5. The voltageconverter according to claim 4, wherein the output switch unit driverseach include a current sense amplifier, a current sense resistorconnected between a positive input of the current sense amplifier and anoutput inductor node, and a current sense capacitor connected betweenthe positive input and a negative input of the current sense amplifier,with the output inductor also being connected to the negative input ofthe current sense amplifier, wherein the current sense resistor and thecurrent sense capacitor are selected such that a resulting time constantequals a time constant of the inductance of the output inductor and a DCresistance thereof, whereby each of the output phase arrangements sensesthe current provided to the load thereby without interfering with thecurrent provided to the load.
 6. The voltage converter according toclaim 3, wherein the output switch units and the output circuit areconfigured to provide buck converter topology.
 7. The voltage converteraccording to claim 3, wherein the output switch unit drivers are furtheroperative to turn off the high-side switch and the low-side switch inresponse to at least one of a request for a lower desired voltage and adecrease in a current demand of the load.
 8. The voltage converteraccording to claim 7, wherein the request for the lower desired voltageand the decrease in the current demand of the load are determined fromthe PWM control signal.
 9. The voltage converter according to claim 3,wherein: the master control unit provides a PWM control signal to eachof the output switch unit drivers, and the output switch unit driversare each responsive to a total current provided to the load to modifythe PWM control signal, such that the output voltage is reducedproportionally to an increase in the total current provided to the load.10. The voltage converter according to claim 3, wherein the outputswitch unit drivers are further operative to turn off the high-sideswitch and the low-side switch in response to at least one of a requestfor a lower desired voltage and a decrease in a current demand of theload.
 11. The voltage converter according to claim 3, wherein: themaster control unit provides a PWM control signal to each of the outputswitch unit drivers, and the output switch unit drivers are furtheroperative to modify the PWM control signal such that the phase outputunits switch off the high-side and low-side switches of the respectiveoutput switch units in response to a request for a lower desired outputvoltage.
 12. The voltage converter according to claim 1, wherein theoutput switch units and the output circuit are configured to providebuck converter topology.
 13. The voltage converter according to claim 1,wherein: the master control unit provides phase timing signals, and aPWM control signal to each of the output switch unit drivers, and theoutput switch unit drivers are respectively responsive to one of thephase timing signals to turn on a switch in the associated output switchunit, and responsive to the PWM control signal to turn off the switch.14. The voltage converter according to claim 13, wherein the mastercontrol unit provides the phase timing signals and the PWM controlsignal over a common control bus to all of the output switch unitdrivers.
 15. The voltage converter according to claim 1, wherein theoutput switch unit drivers each include a phase error detection unitoperative to provide a phase error signal according to a differencebetween the phase output current for the respective output switch unitand a signal representing an average phase output current.
 16. Thevoltage converter according to claim 1, wherein the switch unit driverseach include a phase error detection circuit responsive to a conditionin which the respective phase is incapable of providing a phase outputcurrent to match an average inductor current of all of the phases todeactivate a defective phase and/or enable a back-up phase.